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FEATURES

LTC3419 Dual Monolithic 600mA Synchronous Step-Down Regulator DESCRIPTION
The LTC(R)3419 is a dual, 2.25MHz, constant-frequency, synchronous step-down DC/DC converter in a tiny 3mm x 3mm DFN package. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Low output voltages are supported with the 0.6V feedback reference voltage. Each regulator can supply 600mA output current. The input voltage range is 2.5V to 5.5V, making it ideal for Li-Ion and USB powered applications. Supply current during operation is only 35A and drops to <1A in shutdown. A user-selectable mode input allows the user to trade off between high efficiency Burst Mode operation and pulse-skipping mode. An internally set 2.25MHz switching frequency allows the use of tiny surface mount inductors and capacitors. Internal soft-start reduces inrush current during start-up. Both outputs are internally compensated to work with ceramic output capacitors. The LTC3419 is available in a low profile (0.75mm) 3mm x 3mm DFN package. The LTC3419 is also available in a fixed output voltage configuration selected via internal resistor dividers (see Table 2).
, LT, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6127815, 6304066, 6498466, 6580258, 6611131.

High Efficiency Dual Step-Down Outputs: Up to 96% 600mA Current per Channel at VIN = 3V Only 35A Quiescent Current During Operation (Both Channels) 2.25MHz Constant-Frequency Operation 2.5V to 5.5V Input Voltage Range Low Dropout Operation: 100% Duty Cycle No Schottky Diodes Required Internally Compensated for All Ceramic Capacitors Independent Internal Soft-Start for Each Channel Available in Fixed Output Versions Current Mode Operation for Excellent Line and Load Transient Response 0.6V Reference Allows Low Output Voltages User-Selectable Burst Mode(R) Operation Short-Circuit Protected Ultralow Shutdown Current: IQ < 1A Available in Small MSOP or 3mm x 3mm DFN-8 Packages
APPLICATIONS

Cellular Telephones Digital Still Cameras Wireless and DSL Modems Portable Media Players PDAs/Palmtop PCs
TYPICAL APPLICATION
Dual Monolithic Buck Regulator in 8-Lead 3 x 3 DFN
VIN 2.5V TO 5.5V 10F RUN2 VIN RUN1 MODE VOUT2 1.8V AT 600mA 3.3H 22pF LTC3419 SW2 SW1 22pF 3.3H VOUT1 2.5V AT 600mA
Efficiency and Power Loss vs Output Current
100 90 80 70 EFFICIENCY (%) 60 50 40 30 0.01 0.1 1 POWER LOSS (W) VIN = 3.6V 10
VFB2 10F 118k 59k
GND
VFB1 59k 187k 10F
3419 TA01
20 10 0 0.1
VOUT = 1.2V VOUT = 1.8V VOUT = 2.5V 1 10 100 OUTPUT CURRENT (mA)
0.001
0.0001 1000
3419 TA01b
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LTC3419 ABSOLUTE MAXIMUM RATINGS
(Note 1)
Input Supply Voltage (VIN) ............................. -0.3 to 6V VFB1, VFB2 ........................................ -0.3V to VIN + 0.3V RUN1, RUN2, MODE ........................ -0.3V to VIN + 0.3V SW1, SW2 ....................................... -0.3V to VIN + 0.3V P-Channel SW Source Current (DC) (Note 2).......800mA N-Channel SW Source Current (DC) (Note 2) ......800mA
Peak SW Source and Sink Current (Note 2) .............1.3A Operating Temperature Range (Note 3) .......-40 to 85C Junction Temperature (Note 6) ............................. 125C Storage Temperature Range...................-65C to 125C Lead Temperature (Soldering, 10 sec) MSOP Package ................................................. 300C
PACKAGE/ORDER INFORMATION
TOP VIEW VFB1 1 RUN1 2 MODE 3 SW1 4 9 8 7 6 5 VFB2 RUN2 SW2 VIN TOP VIEW VFB1 RUN1 MODE SW1 GND 1 2 3 4 5 10 9 8 7 6 VFB2 RUN2 SW2 VIN GND
DD PACKAGE 8-LEAD (3mm x 3mm) PLASTIC DFN TJMAX = 125C, JA = 40C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
MS PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 125C, JA = 120C/W
ORDER PART NUMBER LTC3419EDD LTC3419EDD-1
DD PART MARKING LCQJ LCWW
ORDER PART NUMBER LTC3419EMS LTC3419EMS-1
MS PART MARKING LTCQK LTCWX
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
SYMBOL VIN VUV IFB VFBREG1 PARAMETER VIN Operating Voltage VIN Undervoltage Lockout Feedback Pin Input Current
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C, VIN = 3.6V, unless otherwise noted.
CONDITIONS
MIN 2.5

TYP 2.1 3
MAX 5.5 2.5 30 5 0.610 0.612 1.606 0.610 0.612 1.836 0.5
UNITS V V nA A V V V V V V %/V
VIN Low to High LTC3419 LTC3419-1
Regulated Feedback Voltage (Channel 1) LTC3419, 0C < TA < 85C LTC3419, -40C < TA < 85C LTC3419-1, -40C < TA < 85C Regulated Feedback Voltage (Channel 2) LTC3419, 0C < TA < 85C LTC3419, -40C < TA < 85C LTC3419-1, -40C < TA < 85C Reference Voltage Line Regulation VIN = 2.5V to 5.5V (Note 7)
0.590 0.588 1.544 0.590 0.588 1.764
0.600 0.600 1.575 0.600 0.600 1.8 0.3
VFBREG2
VLINE REG
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LTC3419 ELECTRICAL CHARACTERISTICS
SYMBOL VLOAD REG IS PARAMETER Output Voltage Load Regulation Input DC Supply Current Active Mode (Note 4) Sleep Mode Shutdown Oscillator Frequency Peak Switch Current Limit Channel 1 (600mA) Channel 2 (600mA) Channel 1 (Note 5) Top Switch On-Resistance Bottom Switch On-Resistance Channel 2 (Note 5) Top Switch On-Resistance Bottom Switch On-Resistance Switch Leakage Current Soft-Start Time RUN Threshold High RUN Leakage Current MODE Threshold High MODE Leakage Current Output Ripple in Burst Mode Operation VOUT = 1.5V, COUT = 10F
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C, VIN = 3.6V, unless otherwise noted.
CONDITIONS ILOAD = 0mA to 600mA (Note 7) VFB1 = VFB2 = 0.95 x VFBREG VFB1 = VFB2 = 1.05 x VFBREG , VIN = 5.5V RUN1 = RUN2 = 0V, VIN = 5.5V VFB = VFBREG VIN = 3V, VFB < VFBREG , Duty Cycle < 35% 900 900 VIN = 3.6V, ISW = 100mA VIN = 3.6V, ISW = 100mA VIN = 3.6V, ISW = 100mA VIN = 3.6V, ISW = 100mA VIN = 5V, VRUN = 0V VFB from 10% to 90% Full Scale

MIN
TYP 0.5 500 35 0.1
MAX
UNITS %
700 60 1 2.7
A A A MHz mA mA
fOSC ILIM
1.8
2.25 1200 1200 0.4 0.4 0.4 0.4 0.01
RDS(ON)
0.6 0.6 0.6 0.6 1 1.3 1.2 1 1.2 1
A ms V A V A mVP-P
ISW(LKG) tSOFTSTART VRUN IRUN VMODE IMODE VBURST
0.1 0.4 0.4
0.95 1 0.01 1 0.01 20
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Guaranteed by long term current density limitations. Note 3: The LTC3419E and LTC3419E-1 are guaranteed to meet specified performance from 0C to 85C. Specifications over the -40C and 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency.
Note 5: The DFN switch on-resistance is guaranteed by correlation to wafer level measurements. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 7: The converter is tested in a proprietary test mode that connects the output of the error amplifier to the SW pin, which is connected to an external servo loop.
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LTC3419 TYPICAL PERFORMANCE CHARACTERISTICS
Burst Mode Operation
SW 2V/DIV VOUT 50mV/DIV AC-COUPLED IL 100mA/DIV
3419 G01
TA = 25C, VIN = 3.6V, unless otherwise noted.
Pulse Skip Mode Operation
100 SW 2V/DIV VOUT 50mV/DIV AC-COUPLED IL 100mA/DIV EFFICIENCY (%) 90 80 70 60 50 5s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 5mA
3419 G02
Efficiency vs Input Voltage
IOUT = 100mA
IOUT = 1mA IOUT = 600mA IOUT = 10mA
IOUT = 0.1mA 40 VOUT = 1.8V 30 3.0 3.5 2.5
2s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 25mA
4.0 VIN (V)
4.5
5.0
5.5
3419 G03
Reference Voltage vs Temperature
1.5 1.0 VFB (% ERROR) 0.5 0 -0.5 -1.0 -1.5 -50 FREQUENCY (MHz) 2.6 2.5
Oscillator Frequency vs Temperature
60
Supply Current vs Temperature
ILOAD = 0A RUN1 = RUN2 = VIN
50 2.3 2.2 VIN = 2.7V 2.1 2.0 1.9 -25 0 50 25 75 TEMPERATURE (C) 100 125 1.8 -50 -25 25 75 0 50 TEMPERATURE (C) 100 125 10 -50 -25 25 0 50 TEMPERATURE (C) 75 100
3419 G06
SUPPLY CURRENT (A)
2.4
VIN = 4.2V VIN = 3.6V
40 VIN = 5.5V 30 VIN = 2.7V 20
3419 G04
3419 G05
Switch Leakage vs Input Voltage
3.0 2.5 LEAKAGE CURRENT (nA) 2.0 RDS(ON) () MAIN SWITCH 1.5 1.0 0.5 0 2.5 SYNCHRONOUS SWITCH 0.50 0.45 0.40
Switch On-Resistance vs Input Voltage
0.6
Switch On-Resistance vs Temperature
0.5
MAIN SWITCH
RDS(ON) ()
MAIN SWITCH 0.35 0.30 0.25 0.20 2.5
0.4
0.3 SYNCHRONOUS SWITCH 0.2 VIN = 2.7V VIN = 3.6V VIN = 4.2V -25 25 75 0 50 TEMPERATURE (C) 100 125
SYNCHRONOUS SWITCH
3.0
3.5
4.0 VIN (V)
4.5
5.0
5.5
3419 G07
3.0
3.5
4.0 4.5 VIN (V)
5.0
5.5
6.0
0.1 -50
3419 G08
3419 G09
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LTC3419 TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current
100 90 80 70 EFFICIENCY (%) EFFICIENCY (%) 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
3419 G10
TA = 25C, VIN = 3.6V, unless otherwise noted.
Efficiency vs Load Current
100 90 80 70 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
3419 G11
Efficiency vs Load Current
100 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
3419 G12
VOUT = 1.2V 0 0.1 10 100 1 OUTPUT CURRENT (mA)
VOUT = 1.8V 0 0.1 10 100 1 OUTPUT CURRENT (mA)
VOUT = 2.5V 0 0.1 10 100 1 OUTPUT CURRENT (mA)
Efficiency vs Load Current
100 90 80 VOUT ERROR (%) 70 EFFICIENCY (%) 60 50 40 30 20 10 VOUT = 1.8V 0 0.1 10 100 1 OUTPUT CURRENT (mA) 1000
3419 G13
Load Regulation
3.0 2.5 2.0 VOUT ERROR (%) VOUT = 1.2V VOUT = 1.8V VOUT = 2.5V 2.0 1.5 1.0 0.5 0 -0.5 -1.0
Load Regulation
VOUT = 1.8V
Burst Mode OPERATION
PULSE SKIP MODE
1.5 Burst Mode OPERATION 1.0 0.5 0 -0.5 -1.0 0 100 200 400 300 LOAD CURRENT (mA) 500 600
3419 G14
Burst Mode OPERATION PULSE SKIP MODE 0 100 300 200 400 LOAD CURRENT (mA) 500 600
3419 G15
Line Regulation
0.6 0.4 VOUT ERROR (%) 0.2 0 -0.2 -0.4 -0.6 2.5 IL 500mA/DIV VOUT = 1.8V ILOAD = 100mA RUN 2V/DIV
Start-Up from Shutdown
RUN 2V/DIV
Start-Up from Shutdown
VOUT 1V/DIV
VOUT 1V/DIV ILOAD 500mA/DIV 250s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A
3419 G17
250s/DIV VIN = 3.6V VOUT = 1.8V RLOAD = 3
3419 G18
3.0
3.5
4.0 VIN (V)
4.5
5.0
5.5
3419 G16
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LTC3419 TYPICAL PERFORMANCE CHARACTERISTICS
Load Step
VOUT 100mV/DIV AC-COUPLED IL 500mA/DIV ILOAD 500mA/DIV
3419 G19
TA = 25C, VIN = 3.6V, unless otherwise noted.
Load Step
VOUT 100mV/DIV AC-COUPLED IL 500mA/DIV ILOAD 500mA/DIV 20s/DIV 20s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 40mA TO 600mA
3419 G20
Load Step
VOUT 100mV/DIV AC-COUPLED IL 500mA/DIV ILOAD 500mA/DIV 20s/DIV VIN = 3.6V VOUT = 1.2V ILOAD = 40mA TO 600mA
3419 G21
VIN = 3.6V VOUT = 1.8V ILOAD = 0A TO 600mA
PIN FUNCTIONS
(DD/MS)
VFB1 (Pin 1/Pin 1): Regulator 1 Output Feedback. Receives the feedback voltage from the external resistive divider across the regulator 1 output. Nominal voltage for this pin is 0.6V. RUN1 (Pin 2/Pin 2): Regulator 1 Enable. Forcing this pin to VIN enables regulator 1, while forcing it to GND causes regulator 1 to shut down. MODE (Pin 3/Pin 3): Mode Select Input. To select pulseskipping mode, tie to VIN. Grounding this pin selects Burst Mode operation. Do not leave this pin floating. SW1 (Pin 4/Pin 4): Regulator 1 Switch Node Connection to the Inductor. This pin swings from VIN to GND. VIN (Pin 5/Pin 7): Main Power Supply. Must be closely de-coupled to GND.
SW2 (Pin 6/Pin 8): Regulator 2 Switch Node Connection to the Inductor. This pin swings from VIN to GND. RUN2 (Pin 7/Pin 9): Regulator 2 Enable. Forcing this pin to VIN enables regulator 2, while forcing it to GND causes regulator 2 to shut down. VFB2 (Pin 8/Pin 10): Regulator 2 Output Feedback. Receives the feedback voltage from the external resistive divider across the regulator 2 output. Nominal voltage for this pin is 0.6V. Exposed Pad (Pin 9/NA): Ground. The Exposed Pad must be soldered to PCB for optimal thermal performance. GND (NA/Pins 5, 6): Ground. Connect to the (-) terminal of COUT, and the (-) terminal of CIN. Pin 5 of the MS package must be soldered to the PC board for optimal thermal performance.
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LTC3419 FUNCTIONAL DIAGRAM
REGULATOR 1 MODE 3 BURST CLAMP SLOPE COMP 5 VIN
VFB1
1
-
EA 0.6V
-
ITH VSLEEP SLEEP
-
ICOMP
+
+
+
BURST S Q RS LATCH R Q SWITCHING LOGIC AND BLANKING CIRCUIT
SOFT-START
ANTI SHOOTTHRU 4 SW1
IRCMP SHUTDOWN
RUN1 RUN2
2 0.6V REF 7 OSC OSC
SLEEP2
SLEEP1
VFB2
8
REGULATOR 2 (IDENTICAL TO REGULATOR 1)
3419 FD
-
+
9 GND 6 SW2
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LTC3419 OPERATION
The LTC3419 uses a constant-frequency, current mode architecture. The operating frequency is set at 2.25MHz. Both channels share the same clock and run in-phase. The output voltage is set by an external resistor divider returned to the VFB pins. An error amplifier compares the divided output voltage with a reference voltage of 0.6V and regulates the peak inductor current accordingly. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle when the VFB voltage is below the reference voltage. The current into the inductor and the load increases until the peak inductor current (controlled by ITH) is reached. The RS latch turns off the synchronous switch and energy stored in the inductor is discharged through the bottom switch (N-channel MOSFET) into the load until the next clock cycle begins, or until the inductor current begins to reverse (sensed by the IRCMP comparator). The peak inductor current is controlled by the internally compensated ITH voltage, which is the output of the error amplifier. This amplifier regulates the VFB pin to the internal 0.6V reference by adjusting the peak inductor current accordingly. Light Load Operation There are two modes to control the LTC3419 at light load currents: Burst Mode operation and pulse-skipping mode. Both automatically transition from continuous operation to the selected mode when the load current is low. To optimize efficiency, Burst Mode operation can be selected by grounding the MODE pin. When the load is relatively light, the peak inductor current (as set by ITH) remains fixed at approximately 60mA and the PMOS switch operates intermittently based on load demand. By running cycles periodically, the switching losses are minimized. The duration of each burst event can range from a few cycles at light load to almost continuous cycling with short sleep intervals at moderate loads. During the sleep intervals, the load current is being supplied solely from the output capacitor. As the output voltage droops, the error amplifier output rises above the sleep threshold, signaling the burst comparator to trip and turn the top MOSFET on. This cycle repeats at a rate that is dependent on load demand. For applications where low ripple voltage and constantfrequency operation is a higher priority than light load efficiency, pulse-skipping mode can be used by connecting the MODE pin to VIN. In this mode, the peak inductor current is not fixed, which allows the LTC3419 to switch at a constant-frequency down to very low currents, where it will begin skipping pulses. Dropout Operation When the input supply voltage decreases toward the output voltage the duty cycle increases to 100%, which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. An important design consideration is that the RDS(ON) of the P-channel switch increases with decreasing input supply voltage (see Typical Performance Characteristics). Therefore, the user should calculate the worst-case power dissipation when the LTC3419 is used at 100% duty cycle with low input voltage (see Thermal Considerations in the Applications Information section). Soft-Start In order to minimize the inrush current on the input bypass capacitor, the LTC3419 slowly ramps up the output voltage during start-up. Whenever the RUN1 or RUN2 pin is pulled high, the corresponding output will ramp from zero to full-scale over a time period of approximately 750s. This prevents the LTC3419 from having to quickly charge the output capacitor and thus supplying an excessive amount of instantaneous current. Short-Circuit Protection When either regulator output is shorted to ground, the corresponding internal N-channel switch is forced on for a longer time period for each cycle in order to allow the inductor to discharge, thus preventing inductor current runaway. This technique has the effect of decreasing switching frequency. Once the short is removed, normal operation resumes and the regulator output will return to its nominal voltage.
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LTC3419 APPLICATIONS INFORMATION
A general LTC3419 application circuit is shown in Figure 1. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance and increases with higher VIN or VOUT : V IL = OUT fO * L V * 1 - OUT VIN (1) or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price versus size requirements, and any radiated field/EMI requirements, than on what the LTC3419 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3419 applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER Taiyo Yuden PART NUMBER CB2016T2R2M CB2012T2R2M CB2016T3R3M ELT5KT4R7M CDRH2D18/LD LQH32CN4R7M23 NR30102R2M NR30104R7M FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D VLF3010AT4R7MR70 VLF3010AT3R3MR87 VLF3010AT2R2M1R0 MAX DC VALUE CURRENT 2.2H 2.2H 3.3H 4.7H 4.7H 4.7H 2.2H 4.7H 4.7H 3.3H 2.2H 4.7H 3.3H 2.2H 510mA 530mA 410mA 950mA 630mA 450mA 1100mA 750mA 1100mA 1200mA 1300mA 700mA 870mA 1000mA DCR HEIGHT
0.13 1.6mm 0.33 1.25mm 0.27 1.6mm 0.2 0.086 0.2 0.1 0.19 0.11 0.1 0.08 0.28 0.17 0.12 1.2mm 2mm 2mm 1mm 1mm 1mm 1mm 1mm 1mm 1mm 1mm
Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is 40% of the maximum output load current. So, for a 600mA regulator, IL = 240mA (40% of 600mA). The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the peak inductor current falls below a level set by the internal burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. Furthermore, lower inductance values will cause the bursts to occur with increased frequency. Inductor Core Selection Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid
VIN 2.5V TO 5.5V C1 RUN2 VIN RUN1 MODE L2 CF2 LTC3419 SW2 SW1 CF1 L1 VOUT1
Panasonic Sumida Murata Taiyo Yuden FDK
TDK
Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT / VIN . To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT ( VIN - VOUT ) VIN
VOUT2
VFB2 COUT2 R4 R3
GND
VFB1 R1 R2 COUT1
3419 F01
Where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - IL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant
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Figure 1. LTC3419 General Schematic
9
LTC3419 APPLICATIONS INFORMATION
deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling when not using an all-ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by: 1 VOUT IL ESR + 8 fOCOUT where fO = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3419 control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. For more information, see Application Note 88. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Setting the Output Voltage The LTC3419 regulates the VFB1 and VFB2 pins to 0.6V during regulation. Thus, the output voltage is set by a resistive divider according to the following formula: R2 VOUT = 0 . 6 V 1 + R1 (2)
Keeping the current small (< 10A) in these resistors maximizes efficiency, but making it too small may allow stray capacitance to cause noise problems or reduce the phase margin of the error amp loop. To improve the frequency response of the main control loop, a feedback capacitor (CF) may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Fixed output versions of the LTC3419 (e.g. LTC3419-1) include an internal resistive divider, eliminating the need for external resistors. The resistor divider is chosen such that the VFB input current is approximately 3A. For these versions the VFB pin should be connected directly to VOUT. Table 2 lists the fixed output voltages available for the LTC3419.
Table 2. Fixed Output Voltage Versions
PART NUMBER LTC3419 LTC3419-1 VOUT1 Adjustable 1.575V VOUT2 Adjustable 1.8V
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LTC3419 APPLICATIONS INFORMATION
Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD * ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine the phase margin. In addition, feedback capacitors (CF1 and CF2) can be added to improve the high frequency response, as shown in Figure 1. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>1F) input capacitors. The discharged input capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: % Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four sources usually account for the losses in LTC3419 circuits: 1) VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other system losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 2. The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3. I2R losses are calculated from the DC resistances of the internal switches, RSW , and external inductor, RL. In continuous mode, the average output current flows through inductor L, but is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP) * (DC) + (RDS(ON)BOT) * (1- DC)
Hot Swap is a trademark of Linear Technology Corporation.
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11
LTC3419 APPLICATIONS INFORMATION
The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2 * (RSW + RL) 4. Other "hidden" losses, such as copper trace and internal battery resistances, can account for additional efficiency degradations in portable systems. It is very important to include these "system" level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses, including diode conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. Thermal Considerations In a majority of applications, the LTC3419 does not dissipate much heat due to its high efficiency. In the unlikely event that the junction temperature somehow reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. The goal of the following thermal analysis is to determine whether the power dissipated causes enough temperature rise to exceed the maximum junction temperature (125C) of the part. The temperature rise is given by: TRISE = PD * JA Where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TRISE + TAMBIENT As a worst-case example, consider the case when the LTC3419 is in dropout on both channels at an input voltage of 2.7V with a load current of 600mA and an ambient temperature of 70C. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) of the main switch is 0.6. Therefore, power dissipated by each channel is: PD = IOUT2 * RDS(ON) = 216mV Given that the thermal resistance of a properly soldered DFN package is approximately 40C/W, the junction temperature of an LTC3419 device operating in a 70C ambient temperature is approximately: TJ = (2 * 0.216W * 40C/W) + 70C = 87.3C which is well below the absolute maximum junction temperature of 125C. PC Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3419. These items are also illustrated graphically in the layout diagrams of Figures 2 and 3. Check the following in your layout: 1. Does the capacitor CIN connect to the power VIN (Pin 5) and GND (Pin 9) as closely as possible? This capacitor provides the AC current of the internal power MOSFETs and their drivers. 2. Are the respective COUT and L closely connected? The (-) plate of COUT returns current to GND and the (-) plate of CIN. 3. The resistor divider, R1 and R2, must be connected between the (+) plate of COUT1 and a ground sense line terminated near GND (Pin 9). The feedback signals VFB1 and VFB2 should be routed away from noisy components and traces, such as the SW lines (Pins 4 and 6), and their trace length should be minimized. 4. Keep sensitive components away from the SW pins, if possible. The input capacitor CIN and the resistors R1, R2, R3 and R4 should be routed away from the SW traces and the inductors. 5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the GND pin at a single point. These ground traces should not share the high current path of CIN or COUT. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to VIN or GND.
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12
LTC3419 APPLICATIONS INFORMATION
VIN 2.5V TO 5.5V C1 RUN2 VIN RUN1 MODE LTC3419 SW2 CF2 SW1 CF1
L2 VOUT2
L1 VOUT1
VFB2 COUT2 R4 R3
GND
VFB1 R1 R2 COUT1
3419 F02
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 2. LTC3419 Layout Diagram (See Board Layout Checklist)
CF1 R2 VOUT1 COUT1 VIA TO VIN VFB1 RUN1 MODE SW1 VIA TO GND GND CIN VFB2 R1 R3
CF2 R4 VOUT2 COUT2
L1
RUN2 SW2 VIN
L2
3419 F03
Figure 3. LTC3419 Suggested Layout
Design Example As a design example, consider using the LTC3419 in a portable application with a Li-Ion battery. The battery provides a VIN ranging from 2.8V to 4.2V. The load on each channel requires a maximum of 600mA in active mode and 2mA in standby mode. The output voltages are VOUT1 = 2.5V and VOUT2 = 1.8V. Start with channel 1. First, calculate the inductor value for about 40% ripple current (240mA in this example) at maximum VIN. Using a derivation of Equation 1: L1 = 2 . 5V 2 . 5V * 1- = 1 . 8 7H 2 . 25MHz * (240mA) 4 . 2V
A 10F ceramic capacitor should be more than sufficient for this output capacitor. As for the input capacitor, a typical value of CIN = 10F should suffice, as the source impedance of a Li-Ion battery is very low. The feedback resistors program the output voltage. To maintain high efficiency at light loads, the current in these resistors should be kept small. Choosing 10A with the 0.6V feedback voltage makes R1~60k. A close standard 1% resistor is 59k. Using Equation 2. V R2 = OUT - 1 * R1 = 187k 0.6 An optional 22pF feedback capacitor (CF1) may be used to improve transient response.
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For the inductor, use the closest standard value of 2.2H.
13
LTC3419 APPLICATIONS INFORMATION
Using the same analysis for channel 2 (VOUT2 = 1.8V), the results are:
EFFICIENCY (%) 100 90 80 70 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
L2 = 1.9H R3 = 59k R4 = 118k CF2 = 22pF Figure 4 shows the complete schematic for this example, along with the efficiency curve and transient response.
VIN 2.5V TO 5.5V
VOUT = 1.8V 0 0.1 1 10 100 OUTPUT CURRENT (mA)
100 90
C1 10F L2 2.2H CF2, 22pF
VOUT2 1.8V AT 600mA
LTC3419 SW2 SW1
L1 2.2H CF1, 22pF
EFFICIENCY (%)
RUN2 VIN RUN1 MODE VOUT1 2.5V AT 600mA
80 70 60 50 40 30
COUT2 10F
VFB2 R4 118k R3 59k
VFB1 GND R1 59k R2 187k
COUT1 10F
3419 F04a
20 10 0 0.1 VOUT = 2.5V
VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
3419 F04b
C1, C2, C3: TAIYO YUDEN JMK316BJ106ML
L1, L2: TDK VLF3010AT2R2M1RD
1 10 100 OUTPUT CURRENT (mA)
Figure 4a. Design Example Circuit
Figure 4b. Efficiency vs Output Current
Load Step
VOUT 100mV/DIV AC-COUPLED IL 500mA/DIV ILOAD 500mA/DIV 20s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 40mA TO 600mA
3419 F04c1
Transient Response
VOUT 100mV/DIV AC-COUPLED IL 500mA/DIV ILOAD 500mA/DIV 20s/DIV VIN = 3.6V VOUT = 2.5V ILOAD = 40mA TO 600mA
3419 F04c2
Figure 4c. Transient Response
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14
LTC3419 PACKAGE DESCRIPTION
DD Package 8-Lead Plastic DFN (3mm x 3mm) (Reference LTC DWG # 05-08-1698)
R = 0.115 TYP 5 0.675 0.05 0.38 0.10 8
3.5 0.05 1.65 0.05 2.15 0.05 (2 SIDES)
PACKAGE OUTLINE
3.00 0.10 (4 SIDES)
1.65 0.10 (2 SIDES)
PIN 1 TOP MARK (NOTE 6) 0.200 REF 0.75 0.05 4 0.25 0.05 2.38 0.10 (2 SIDES) 1
(DD) DFN 1203
0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE
MS Package 10-Lead Plastic MSOP (Reference LTC DWG # 05-08-1661)
0.889 0.127 (.035 .005)
3.00 0.102 (.118 .004) (NOTE 3) 10 9 8 7 6
0.497 0.076 (.0196 .003) REF
5.23 (.206) MIN
3.20 - 3.45 (.126 - .136)
4.90 0.152 (.193 .006)
3.00 0.102 (.118 .004) (NOTE 4)
0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT
1.10 (.043) MAX
12345 0.86 (.034) REF
0.254 (.010) GAUGE PLANE
DETAIL "A" 0 - 6 TYP
DETAIL "A" 0.18 (.007) SEATING PLANE 0.53 0.152 (.021 .006)
0.17 - 0.27 (.007 - .011) TYP
0.50 (.0197) BSC
0.127 0.076 (.005 .003)
MSOP (MS) 0603
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3419 TYPICAL APPLICATION
Dual 600mA Buck Converter
VIN 2.5V TO 5.5V C1 10F L2 3.3H CF2, 22pF
1.8V/1.575V Dual 600mA Buck Converter
VIN 2.5V TO 5.5V C1 10F L2 3.3H
RUN2 VIN RUN1 MODE LTC3419 SW2 SW1 CF1, 22pF L1 3.3H VOUT1 2.5V AT 600mA VOUT2 1.8V AT 600mA
RUN2 VIN RUN1 MODE LTC3419-1 SW2 SW1 L1 3.3H VOUT1 1.575V AT 600mA
VOUT2 1.8V AT 600mA
COUT2 10F
VFB2 R4 118k R3 59k
GND
VFB1 R1 59k R2 187k
COUT1 10F
3419 TA02
COUT2 10F
VFB2
GND
VFB1
COUT1 10F
3419 TA03
C1, C2, C3: TAIYO YUDEN JMK316BJ106ML
L1, L2: TDK VLF3010AT3R3M1RD
C1, C2, C3: TAIYO YUDEN JMK316BJ106ML
L1, L2: TDK VLF3010AT3R3M1RD
RELATED PARTS
PART NUMBER LTC3405/LTC3405A LTC3406/LTC3406B LTC3407/LTC3407-2 DESCRIPTION 300mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC Converters 600mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC Converters Dual 600mA/800mA IOUT, 1.5MHz/ 2.25MHz, Synchronous Step-Down DC/DC Converters 600mA IOUT, 1.7MHz/2.6MHz, Synchronous Step-Down DC/DC Converter 300mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converters 1.25A IOUT, 4MHz, Synchronous Step-Down DC/DC Converter 2.5A IOUT 4MHz, Synchronous Step-Down DC/DC Converter 1.2A IOUT 2MHz, Synchronous Buck-Boost DC/DC Converters COMMENTS 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 20A, ISD = <1A, ThinSOTTM Package 96% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 20A, ISD = <1A, ThinSOT Package 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD = <1A, MS10E and DFN Packages 96% Efficiency, VIN(MIN) = 1.6V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 65A, ISD = <1A, DFN Package 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 26A, ISD = <1A, SC70 Package 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD = <1A, MS10 and DFN Packages 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD = <1A, TSSOP-16E Package 95% Efficiency, VIN(MIN) = 2.4V, VIN(MAX) = 5.5V, VOUT(MIN): 2.4V to 5.25V, IQ = 50A, ISD = <1A, DFN Package 95% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MIN): 2V to 5V, IQ = 16A, ISD = <1A, ThinSOT and DFN Packages 95% Efficiency, VIN(MIN) = 2.4V, VIN(MAX) = 5.5V, VOUT(MIN): 2.4V to 5.25V, IQ = 35A, ISD = <1A, MS10 and DFN Packages 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN): 0.6V, IQ = 40A, ISD = <1A, DFN-8 Package 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN): 0.6V, IQ = 40A, ISD = <1A, MS10E and DFN Packages 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN): 0.8V, IQ = 240A, ISD = <1A, DFN Package
LTC3409
LTC3410/LTC3410B LTC3411 LTC3412 LTC3441/LTC3442, LTC3443
LTC3531/LTC3531-3/ 200mA IOUT, 1.5MHz, Synchronous Buck-Boost DC/DC Converter LTC3531-3.3 LTC3532 LTC3547/LTC3547B 500mA IOUT, 2MHz, Synchronous Buck-Boost DC/DC Converter Dual 300mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converters
LTC3548/LTC3548-1/ Dual 400mA and 800mA IOUT, LTC3548-2 2.25MHz, Synchronous Step-Down DC/DC Converters LTC3561 1.25A IOUT, 4MHz, Synchronous Step-Down DC/DC Converter
ThinSOTTM is a trademark of Linear Technology Corporation.
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16 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0407 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2007

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